1、 河南理工大学毕业设计1AC Voltage and Current Sensorless Control ofThree-Phase PWM RectifiersDong-Choon Lee, Member, IEEE, and Dae-Sik Lim1 THREE-PHASE PWM RECTIFIERSA System ModelingFig.1 shows the power circuit of the three-phase PWM rectifier. The voltage equations are given by(1)0a ab bc ccRpLRpLeiv Fig. 1
2、. Three-phase PWM rectifier without ac-side sensors.where , and are the source voltage, the line current, and the rectifier input voltage, respectively and are the input resistance and the input inductance, respectively. When the peak line voltage , angular frequency , and initial phase angle are gi
3、ven, assuming a balanced three-phase system, the source phase voltage is expressed as(2)cos2()3csabcEeWhere(3)0tA transformation matrix based on the estimated phase angle ,which transforms three-phase variables into a synchronous dq reference frame, is河南理工大学毕业设计2(4)232233cos()cos()23iniinMMC Transfo
4、rming (1) into the reference frame using (4)(5)qc qccMd deivRpLwhere p is a differential operator and . Expressing (5) in a vector notation(6)MeRiLJipvwhere, , , (7)qcdeqcdiqcdv01JTaking a transformation of (2) by using (4)(8)cosinEeWhere(9)MExpressing (6) and (8) in a discrete domain, by approximat
5、ing the derivative term in (6) by a forward difference 9, respectively, (10)(1)()(1)MekRiLJikLvT(11) cos()(1)in1EkekWhere T is the sampling period. 河南理工大学毕业设计3Fig. 2. Overall control block diagram.B System ControlThe PI controllers are used to regulate the dc output voltage and the ac input current.
6、 For decoupling current control, the cross-coupling terms are compensated in a feed forward-type and the source voltage is also compensated as a disturbance. For transient responses without overshoot, the anti-windup technique is employed 10. The overall control block diagram eliminating the source
7、voltage and line current sensors is shown in Fig. 2. The estimation algorithm of source voltages and line currents is described in the following sections.2 PREDICTIVE CURRENT ESTIMATIONThe currents of and can not be calculated instantly since the calculation ()aIk()cItime of the DSP is required. To
8、eliminate the delay effect, a state observer can be used. In addition, the state observer provides the filtering effects for the estimated variable.Expressing (5) in a state-space form,(12)xABu(13)yCwhere, , ,RLA10LB1,qcdixqcdevuAnd y is the output. Transforming (12) and (13) into a discrete domain,
9、 respectively,河南理工大学毕业设计4(14)(1)()()XkFkGU(15)YHwhere,1RTLF0TLGThen, the observer equation adding an error correction term to is given by(16)AA(1)()()()XkkUKYkWhere K is the observer gain matrix and “ ” means the estimated quantity, andis the state variable estimated ahead one sampling period. Subtr
10、acting (15) from A(1)Xk(16), the error dynamic equation of the observer is expressed as(17)(1)()r rekFKCekwhere . Here, it is assumed that the model parameters match well A()rekXwith the real ones. Fig. 3 shows the block diagram of the closed-loop state observer.The state variable error depends only
11、 on the initial error and is independent of the input. For (17) to converge to the zero state, the roots of the characteristic equation of (17) should be located within the unit circle. Fig. 3. Closed-loop state observer. 河南理工大学毕业设计5Fig. 4. Short pulse region. 3 EXPERIMENTS AND DISCUSSIONSA. System
12、Hardware ConfigurationFig.5 shows the system hardware configuration.The source voltage is a three-phase,110V.The input resistance and inductance are 0.06 and 3.3 mH, respectively. The dc link capacitance is 2350F and the switching frequency of the PWM rectifier is 3.5 kHz.Fig. 5. System hardware con
13、figuration.河南理工大学毕业设计6Fig. 6. Dc link currents and corresponding phase currents (in sector V ).The TMS320C31 DSP chip operating at 33.3 MHz is used as a main processor and two 12-b A/D converters are used. One of them is dedicated for detecting the dc link current and the other is used for measuring
14、 the dc output voltage and the source voltages and currents, where ac side quantities are just measured for performance comparison.One of two internal timers in the DSP is employed to decide the PWM control period and the other is used to determine the dc link current interrupt. Considering the rect
15、ifier blanking time of 3.5 s, A/D conversion time of 2.6 s, and the other signal delay time, the minimum pulse width is set to 10 s.A. Experimental Results Fig. 6 shows measured dc link currents and phase currents. In case of sector V of the space vector diagram, the dc link current corresponds to f
16、or the switching state of and for that of . Fig. 7(a) shows the raw dc link current before filtering. It has a lot of ringing components due to the resonance of the leakage inductance and the snubber capacitor. When the dc current is sampled at the end point of the active voltage vectors as shown in
17、 the figure, the measuring error can be reduced.河南理工大学毕业设计7Fig. 7. Sampling of dc link currents.Fig. 8. Estimated source voltage and current at starting. To reduce this error further, the low pass filter should be employed, of which result is shown in Fig. 7(b). The cut-off frequency of the Butterwo
18、rths second-order filter is 112 kHz and its delay time is about 2 sec. Since the ringing frequency is 258 kHz and the switching frequency is 3.5 kHz, the filtered signal without significant delay is acquired.Fig. 8 shows the estimated source voltage and current at starting. With the proposed initial
19、 estimation strategy, the starting operation is well performed. Fig. 9 shows the phaseangle, magnitude, and waveform of the estimated source voltage, which coincide well with measured ones.Fig. 10 shows the source voltage and current waveform at unity power factor. Figs. With the estimated quantitie
20、s for the feedback control, the control performance is satisfactory. The dc voltage variation for load changes will be remarkably decreased if a feedforward control for theload current is added, which is possible without additional cur-rent sensor when the PWM rectifier is combined with the PWM inve
21、rter for ac motor 河南理工大学毕业设计8drives.Fig. 9. Estimated source voltage in steady state.(a) phase angle (b)magnitude (c) waveform.Fig. 10. Source voltage and current waveforms. (a) estimated (b) measured.4 CONCLUSIONSThis paper proposed a novel control scheme of the PWM rectifiers without employing any
22、 ac input voltage and current sensors and with using dc voltage and current sensors only. Reducing the number of the sensors used decreases the system cost as well as improves the system reliability. The phase angle and the magnitude of the source voltage have been estimated by controlling the devia
23、tion between the rectifier current and its model current to be zero. For line current reconstruction, switching states and measured dc link currents 河南理工大学毕业设计9were used. To eliminate the effect of the calculation time delay of the microprocessor, the predictive state observer was used. It was shown
24、 that the estimation algorithm is robust to the parameter variation. The whole algorithm has been implemented for a proto-type 1.5 kVAPWM rectifier system controlled by TMS320C31 DSP. The experimental results have verified that the proposed ac sensor elimination method is feasible.河南理工大学毕业设计10无交流电压、
25、电流传感器的三相 PWM 整流器控制Dong-Choon Lee, Member, IEEE, and Dae-Sik Lim1 三相 PWM 整流器A 系统模型图一所示为三相 PWM 整流器的主电路，电压等式给出如下：(1)0a ab bc ccRpLRpLeiv 图 1 无交流传感器三相 PWM 整流器其中 e，i 和 v 分别是源电压，线电流和整流器的输入电压，R 和 L 分别是输入电阻和输入电感。当已知线电压峰值 E，角频率 和初始相位角 时，假定三相系统是平衡的，则源相位电压可以表达为(2)cos2()3csabcEe其中(3)0t一种基于估计相位角 的变换矩阵，将三相变量变换成一个
26、同步的， 坐标m dq系，这个矩阵是(4)232233cos()cos()23iniinMMC 将（1）式变为 坐标系使用式（4）dq河南理工大学毕业设计11(5)qc qccMd deivRpL其中 p 是一个微分算子且 M将（5）式写成矢量形式(6)eRiLJipv其中, , , (7)qcdeqcdiqcdv01J用式（4）对（2）式进行变换(8)cosinEe其中(9)M通过前向差分来接近微分的限幅，分别将（6）式和（8）式用离散域表示(10)(1)()(1)ekRiLJikLvT(11)cos()(1)in1Ekek其中，T 是采样周期图 2 总的控制模块图河南理工大学毕业设计12B
27、 系统控制PI 控制器是用来调节直流输出电压和交流输入电流的。对于解耦电流控制，交叉耦合项用前馈式补偿，同时，源电压作为扰动的补偿。对于没有过调的暂态响应，引入 anti-windup 技术。消除源电压和线电流传感器的总的控制模块图如图 2 所示。源电压和线电流的估计算法在以后的章节中介绍。2预测电流估计由于 DSP 存在计算时间，所以 和 不能立即计算。为了消除延迟的影()aIkc响，可以使用状态监测器。另外，状态监测器可以对估计变量起到滤波作用。将式（5）用状态空间形式表达为(12)xABu(13)yC其中, ,RLA10LB1C,qcdixqcdevuY 是输出。分别将式（12）和式（1
28、3）分别变换成离散领域(14)(1)()()XkFkGU(15)YHX其中,1RTLF0TLG则加入了误差调整的监测器等式为(16)AA(1)()()()XkFkUKYk河南理工大学毕业设计13其中，k 是监测器增益矩阵， “ ”是指估计量， 是提前一个采样周期A(1)Xk估计的状态变量。用式（15）和减去式（16） ，监测器的动态误差等式表述为(17)(1)()r rekFKCe其中 这里，假设模型参数与真实系统吻合的很好。图 7 所示A()rekX是闭环状态监测器的模块图。状态变量误差仅取决于初始误差，与输入无关。为了使式（17）趋于零状态，典型等式（17）的根应该限制在单位圆内。图 3
29、闭环状态监测器图 4 短脉冲区域3 实验与讨论A 系统硬件构造河南理工大学毕业设计14图 5 系统硬件结构图 6 直流电流和相应相电流 (扇区 5 ).图 5 所示是系统的硬件结构图。源电压是三相 110V。输入电阻和电感分别为0.06 和 3.3mH。直流侧电容为 2350F，PWM 整流器的开关切换频率为 3.5KHZ.使用 TMS320C31 DSP 芯片设定在 33.3MHZ 作为主处理器，同时用到两个 12 位的A/D 转换器：一个用来检测直流侧电流，另一个用来检测直流侧输出电压、源电压和电流。其中直流侧数量只是为了性能比较而测量的。DSP 内部的两个时钟一个是用来决定 PWM 波的
30、控制周期，另一个是用来决定直流侧电流中断。考虑到整流器空白时间 3.5S，A/D 转换时间 2.6S 和其他信号延迟时间，最小脉冲宽度 设定为 10S.minTC、实验结果图 6 所示是测得的直流侧电流和相电流。假设空间矢量图的扇区 V，直流侧电河南理工大学毕业设计15流 对应于 。图 7（a）所示是滤波之前未经处理的直流侧电流。因漏电感和缓dcibi冲电容的共振，会产生噪声成分。如图中所示，当采样动态电压矢量末端的直流电流时，测量误差可以减小。图 7 直流侧电流采样图 8 开始时的估计源电压和电流 为了进一步减少误差，可以使用低通滤波器，结果如图 7（b）所示。Butterworth 的第二
31、顺序滤波器的截止频率是 112KHZ，开关切换频率为 3.5KHZ，所以可以得到没有显著延迟的滤波信号。图 8 所示是开始时估计源电压和电流。使用提出的初始估计策略，开始操作效果很好。图 9 所示是估计源电压的相位角、数值和波形。它们和测量的结果十分吻合。图 10 所是在单位功率因数时源电压和电流波形。当 PWM 整流器与逆变器相连时，在没有额外电流传感器的情况下对交流汽车驾驶来说是可行的。河南理工大学毕业设计16图9稳态时的估计源电压. （a）相位角（b）数值 （c）波形图 10 源电压和电流波形（a ）估计值 （b）测量值4结论这篇文章提出了一种 PWM 整流器新颖的控制方法。这种方法没有使用任何交流输入电压和电流传感器，而仅仅使用直流电压和电流传感器。减少传感器数量可以减少系统费用的同时就提高系统的稳定性。通过控制整流器的电流和它的模型电流的偏差为零，可以估计相位角和源电压的数值。对于线电流重建，使用开关状态和直流侧电流测量。为了消除因微处理器计算时间所带来的延迟影响，使用预测状态监测器。可以看出，估计算法对参数变化是健全的。整个算法已经通过TMS320C31 DSP 作为控制器的 1.5kVAPWM 整流器原型执行。实验结果证明已经证明了提出的消除交流传感器方案的可行性。
